The function of a light receiver is to generate an electrical signal from incident reception light. The detection sensitivity of simple photo diodes is not sufficient in many applications. In an avalanche photo diode (APD), the incident light triggers a controlled avalanche breakthrough (avalanche effect). This multiplies the charge carriers generated by incident photons, and a photo current is produced which is proportional to the light reception level but significantly larger than in a simple PIN diode. In a so-called Geiger mode, the avalanche photo diode is biased above the breakthrough voltage so that even a single charge carrier generated by a single photon can trigger an avalanche, which subsequently recruits all available charge carriers due to the strong field. Hence, the avalanche diode counts individual events like a Geiger counter from which the name is derived. Geiger mode avalanche photo diodes are also called SPAD (Single Photon Avalanche Diode). Another common name is SiPM (Silicon Photo Multiplier), where silicon is a widespread, but not the only used semiconductor and SiPM denotes an entire array of SPADs.
The high radiation sensitivity of SPADs is used in a number of applications. These include medical technology like CT, MRI, or blood tests, optical measuring technology like spectroscopy, distance measurement and three-dimensional imaging, radiation detection in nuclear physics, or uses in telescopes for astrophysics.
Geiger APDs or SPADs thus are very fast, highly sensitive photo diodes on a semiconductor basis. One drawback of the high sensitivity is that not only a measurement photon, but also a weak interference event from ambient light, optical cross talk or dark noise may trigger the avalanche breakthrough. The interference event contributes to the measurement signal with the same relatively strong signal as the received measurement light and is indistinguishable within the signal. The avalanche diode subsequently is insensitive for a dead time of about 5 to 100 ns and is unavailable for further measurements during that time. It is therefore common to interconnect and statistically evaluate multiple SPADs.
In order to actually make use of the signal, it has to be tapped or read out from the SPAD detector element. This is shown in FIG. 1 in a very rough block diagram. A signal detection circuit 150 which outputs the measurement signal in a manner which can be processed in subsequent circuits is connected to the actual detector element 110 having one or usually a plurality of SPADs
Looking at known solutions, in a first step buffered and direct signal detection are to be distinguished. FIG. 2a shows an ideal measurement signal for direct signal detection which enables to analyze the time course of the avalanche breakthroughs and subsequent charging. The sharp falling edges are respective detection events where the bias voltage over the SPAD abruptly breaks down due to incident photons, and then slowly recovers with a time constant in a range of 5 ns as discussed above. The avalanche breakthrough obviously is a very fast event, so that an evaluation of the Geiger current in principle enables fast response times in the range well below 1 ns. However, it is not possible to actually make use of the ideal measurement signal in practice with known signal detection circuits.
In buffered signal detection with digital characteristics, the SPAD signal itself is not passed to the outside, but is evaluated with a threshold in an integrated buffer stage. This is illustrated in FIG. 2b in an example. Although a very usable signal is provided for further processing, there is a loss of information caused by the threshold evaluation. In particular, events occurring too early within the sub-threshold range of the charge are completely lost. This explains why buffered signal detection is less useful for high-frequency signals.
In direct signal detection, the measurement signal should represent the course of the actual signal within the SPAD, i.e. ideally the course according to FIG. 2a. FIGS. 3a-c show some circuits conventionally used for that end. In all cases, the performance is insufficient in particular for high-frequency signals.
In a simple resistor coupling according to FIG. 3a, the signal current or Geiger current is directly converted into a corresponding measurement signal voltage by a connected load resistor (impedance ZSignal). The performance is very limited especially for high-frequency signals and is significantly affected by parasitic capacitances, because the combination of load resistor and parasitic capacitances is a low pass.
An amplifier circuit with a transimpedance amplifier according to FIG. 3b is often used in the prior art. The illustration is simplified in that these amplifiers usually have multiple stages. A feedback resistor (ZSignal) between output of the last amplifier stage and input of the first amplifier stage feeds an attenuated output signal back to the amplifier input and forms a negative feedback for negative gain. Already in a mid-frequency range, i.e. some MHz, the negative feedback of several stages leads to considerable phase shifts because the individual latencies of the amplifier stages add up. Thus, a desired virtual ground at the amplifier input is no longer achieved for higher frequencies. The input resistance is approximately calculated asZin=ZSignal/(1+gain).
With increasing frequency, the gain decreases, and the input resistance increases. The increase of the resistance with frequency corresponds to an inductive behavior. Combined with parasitic input capacitances, a resonant circuit is formed, which may lead to instability and a tendency to oscillate.
A further problem of signal detection with a transimpedance amplifier is the matching of the input signal coming from the SPAD detector and the active part of the amplifier. Signal voltages at the virtual ground which also is the input of the amplifier are minimal. At the same time high input impedances for the amplifier element, e.g. an operational amplifier, are common. As a consequence, only very small signal powers arrive at the active part of the amplifying element, namely, the product of the minimal signal voltage at the virtual ground and the input current being very small due to the input impedance. The major part of the signal current coming from the SPAD detector flows through the feedback resistor and thus is not available for the amplifying element in the first place. This unfavorable coupling results in a poor signal-to-noise-ratio.
A third conventional solution shown in FIG. 3c is based on impedance matching using transformers, in this case high-frequency transformers called balun transformers. Such transformers are only suitable up to one GHz. In practice, even this frequency range is not available, because it is only an ideal value which is not achieved since the circuit impedances deviate from 50 Ohm due to stray inductances and parasitic capacitances. There are also variants of transmission line transformers for higher frequencies up to about 3 GHz and special cases like the Guanella transformer. However, they have other unfavorable properties rendering them unsuitable for optimal signal detection, for example because they cannot be operated as an autotransformer or only allow application in certain impedance ranges which do not match a SPAD detector.
WO 2011/117309 A2 proposes to provide a third electrode at the SPAD detector, in addition to the anode and cathode for applying the bias voltage, as a capacitively coupled output for the Geiger current. This is to prevent delaying the readout by circuit elements of the biasing. However, the document does not deal with the actual signal detection.